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| TK651xx STEP-UP VOLTAGE CONVERTER WITH VOLTAGE MONITOR FEATURES s s s s s s s s s s Guaranteed 0.9 V Operation Very Low Quiescent Current Internal Bandgap Reference High Efficiency MOS Switching Low Output Ripple Microprocessor Reset Output Laser-Trimmed Output Voltage Laser-Trimmed Oscillator Undervoltage Lockout Regulation by Pulse Burst Modulation (PBM) APPLICATIONS s s s s s s s Battery Powered Systems Cellular Telephones Pagers Personal Communications Equipment Portable Instrumentation Portable Consumer Equipment Radio Control Systems DESCRIPTION The TK651xx low power step-up DC-DC converter is designed for portable battery powered systems, capable of operating from a single battery cell down to 0.9 V. The TK651xx provides the power switch and the control circuit for a boost converter. The converter takes a DC input and boosts it up to a regulated 2.7, 3.0 or 3.3 V output . The output voltage is laser-trimmed. A Low Output Indicator detector (LOI) monitors the output voltage and provides an active low microprocessor reset signal whenever the output voltage falls below an internally preset limit. An internal Undervoltage Lockout (UVLO) circuit is utilized to prevent the inductor switch from remaining in the "on" mode when the battery voltage is too low to permit normal operation. Pulse Burst Modulation (PBM) is used to regulate the voltage at the VOUT pin of the IC. PBM is the process in which an oscillator signal is gated or not gated to the switch drive each period. The decision is made just before the start of each cycle and is based on comparing the output voltage to an internally-generated bandgap reference. The decision is latched, so the duty ratio is not modulated within a cycle. The average duty ratio is effectively modulated by the "bursting" and skipping of pulses which can be seen at the SW pin of the IC. Special care should be taken to achieve reliability through the use of Oxide, double Nitride passivation. The TK651xx is available in a miniature 6-pin SOT-23L-6 surface mount package. Customized levels of accuracy in oscillator frequency and output voltage are available. TK651xx VIN GND LOI 20 P GND VOUT SW SW BLOCK DIAGRAM VOUT ORDERING INFORMATION Vref UVLO TK651xxM Tape/Reel Code Voltage Code VIN CONTROL CIRCUIT LOI OSCILLATOR VOLTAGE CODE 27 = 2.7 V 30 = 3.0 V 33 = 3.3 V TAPE/REEL CODE TL: Tape Left GND January 1999 TOKO, Inc. Page 1 TK651xx ABSOLUTE MAXIMUM RATINGS All Pins Except SW and GND .................................... 6 V SW Pin ....................................................................... 9 V Power Dissipation (Note 1) ................................ 400 mW Storage Temperature Range ................... -55 to +150 C Operating Temperature Range ...................-20 to +80 C Junction Temperature ........................................... 150 C TK651xx ELECTRICAL CHARACTERISTICS Over operating temperature range and supply voltage range, unless otherwise specified. SYMBOL fOSC VOUT(REG) VOUT(LOAD) VOUT(LINE) DOSC VLOI PARAMETER Internal Oscillator Frequency Regulation Threshold of VOUT Load Regulation of VOUT(REG) Line Regulation of VOUT(REG) On-time Duty Ratio of Oscillator VOUT During LOI Transition TEST CONDITIONS VIN = 1.3 V, IOUT = 0 mA TA = 25 C VIN = 1.3 V, IOUT = 0 to 4 mA VIN = 0.25 V TA = 25 C TA = 25 C MIN 70 -5% TYP 83 VOUT 0 MAX 102 +3% 50 20 55 +4% UNITS kH z V mV mV % V -20 45 -5% 0 50 0.87 VOUT Note 1: Derate at 0.8 mW/oC for operation above TA = 25 oC ambient temperature, when heat conducting copper foil path is maximized on the printed circuit board. When this is not possible, a derating factor of 1.6 mW/ C must be used. GENERAL CIRCUIT V IN I(V IN ) GND LOI GND 300 k SW V OUT I(V OUT ) I OUT V OUT IB V IN L D C Page 2 January 1999 TOKO, Inc. TK651xx FINAL TEST CIRCUIT VIN LOI 300 k GND GND VOUT IOUT VOUT CS 220 pF D RS + 15 CU 10 F + CD 10 F CN 10 F SW IB VIN L = 95 F ROF Note: Inductor L: Toko A682AE-014 or equivalent Diode D: LL101 Capacitors CN:CU:CD: Panasonic TE series, ECS-TOJY106R 1K Above is the Final Test Circuit through which each of the production parts must pass. In this test circuit, the part is tested against the specification limits in the data sheet (the min. and max. values in the Electrical Characteristics) at room temperature, and is rejected if the tested values are outside the minimum (min.) and maximum (max.) values. The Bench Test Circuits shown on the following pages are the circuits used most of the time to measure the typical (typ.) values in the Electrical Characteristics section, and make the Typical Performance graphs. Note: In measuring the oscillator frequency and the Max IOUT on the bench, the converter was loaded until "no pulse skipping" mode was achieved. January 1999 TOKO, Inc. Page 3 TK651xx TK65127 ELECTRICAL CHARACTERISTICS Over operating temperature range and supply voltage range, unless otherwise specified. SYMBOL VIN IB(Q) I(VIN) I(VOUT) fOSC /T VOUT(REG) VOUT /T VOUT(LOI) VOUT(LOI) RSW(ON) EFF VULV PARAMETER Supply Voltage No Load Battery Current (Note 3) Quiescent Current into VIN Pin Quiescent Current into VOUT Pin Temperature Stability of Oscillator Regulation Threshold of VOUT Temperature Stability of VOUT(REG) VOUT During LOI Transition VOUT(LOI) Threshold Hysteresis On-resistance of SW Pin Converter Efficiency (Notes 2,3) Undervoltage Lockout Voltage TEST CONDITIONS MIN 0.90 TYP MAX 1.60 UNITS V A A A %/ C VIN = 1.3 V, IOUT = 0 mA, T A = 25 C VIN = 1.3 V, IOUT = 0 mA, T A = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, No Pulse Skipping T A = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, TA = 25 C TA = 25 C VOUT 2.4 V VIN = 1.3 V, IOUT = 6 mA, L = 95 H, 3DF Coil TA = 25 C, (Note 4) VIN = 1.1 V, TA = 25 C, L = 95 H, 3DF Coil VIN = 1.3 V, TA = 25 C, L = 95 H, 3DFCoil VIN = 1.1 V, TA = 25 C, L = 39 H, D73 Coil VIN = 1.3 V, TA = 25 C, L = 39 H, D73 Coil 6 8 2.24 2.56 56 12.5 14.5 0.1 2.70 100 2.36 38 1.0 76 0.45 7.6 12.8 15.5 33.6 84 20 23 2.79 V ppm/ C 2.45 V mV % 0.79 V mA mA mA mA IOUT(MAX) Maximum IOUT for Converter (Notes 1,3) Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, RN ECS-TOJY106R 1K BENCH TEST CIRCUIT I(VIN) VIN LOI 300 k Note 1: Note 2: CN 10 F GND GND VOUT Note 3: IND IB VIN L = 95 H CB 10 F CS 220 pF D RS I(VOUT) IOUT VOUT CO 10 F Note 4: Maximum load current depends on inductor value and input voltages. Output ripple depends on filter capacitor values, ESRs and the inductor value. When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Regulation not guaranteed. 1K Page 4 January 1999 TOKO, Inc. TK651xx TYPICAL PERFORMANCE CHARACTERISTICS TK65127 OSCILLATOR FREQUENCY VS. TEMPERATURE 95 OUTPUT REGULATION VOLTAGE VS. TEMPERATURE 2.80 BATTERY CURRENT VS. INPUT VOLTAGE 120 100 VOUT(REG) (V) 90 f OSC (kHz) TA = 25 C NO LOAD 2.75 80 2.70 IB (A) 0 50 100 60 40 2.65 20 85 80 75 -50 0 50 100 2.60 -50 0 0 .5 1 1.5 VIN (V) 2 2.5 3 TEMPERATURE (C) TEMPERATURE (C) OUTPUT VOLTAGE VS. LOAD CURRENT 2.8 L = 95 H TOKO P/N: A682AE-014 (3DF SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 2.8 L = 100 H TOKO P/N: A636CY-101M (D73 SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 2.8 L = 39 H TOKO P/N: A636CY-390M (D73 SERIES) TA = 25 C TA = 25 C TA = 25 C 2.7 2.7 2.7 VIN = 0.9 V VOUT (V) VOUT (V) 2.6 VIN = 0.9 V 1.3 V 2.6 1.3 V VOUT (V) 2.6 VIN = 0.9 V 1.3 V 1.1 V 2.5 1.6 V 2.5 1.1 V 1.6 V 2.5 1.1 V 1.6 V 2.4 1 10 IOUT (mA) 100 2.4 1 10 IOUT (mA) 100 2.4 1 10 IOUT (mA) 100 EFFICIENCY VS. LOAD CURRENT 90 85 80 EFF (%) L = 95 F Toko P/N: A682AE-014 (3DF SERIES) SMALL COIL EFFICIENCY VS. LOAD CURRENT 90 85 80 EFF (%) L = 100 F Toko P/N: 636CY-101M (D73 SERIES) LARGER COIL MAXIMUM OUTPUT CURRENT VS. INDUCTOR VALUE (H) 50 40 IOUT(MAX) (mA) 30 20 1.1 V 10 VIN = 0.9 V 0 0 1.3 V NO PULSE SKIPPING MODE TA = 25 C TA = 25 C TA = 25 C 1.1 V 1.3 V 1.6 V 1.1 V 1.3 V 1.6 V VIN = 0.9 V 75 70 65 60 0.1 1 10 IOUT (mA) 100 75 70 65 60 0.1 1 10 IOUT (mA) VIN = 0.9 V 100 40 80 120 160 INDUCTOR VALUE (H) January 1999 TOKO, Inc. Page 5 TK651xx TK65130 ELECTRICAL CHARACTERISTICS Over operating temperature range and supply voltage range, unless otherwise specified. SYMBOL VIN IB(Q) I(VIN) I(VOUT) fOSC /T VOUT(REG) VOUT /T VOUT(LOI) VOUT(LOI) RSW(ON) EFF VULV PARAMETER Supply Voltage No Load Battery Current (Note 3) Quiescent Current into VIN Pin Quiescent Current into VOUT Pin Temperature Stability of Oscillator Regulation Threshold of VOUT Temperature Stability of VOUT(REG) VOUT During LOI Transition VOUT(LOI) Threshold Hysteresis On-resistance of SW Pin Converter Efficiency (Notes 2,3) Undervoltage Lockout Voltage TEST CONDITIONS MIN 0.90 TYP MAX 2.50 UNITS V A A A %/ C VIN = 1.3 V, IOUT = 0 mA, TA = 25 C VIN = 1.3 V, IOUT = 0 mA, TA = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, No Pulse Skipping TA = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, TA = 25 C TA = 25 C VOUT 2.4 V 2.48 2.85 79 20 22 0.1 3.00 100 2.82 60 1.0 77 0.45 4 6 6.7 10.8 14.0 28.6 111 35 40 3.10 V ppm/ C 2.70 V mV % VIN = 1.3 V, IOUT = 6 mA, L = 95 H, 3DF Coil TA = 25 C, (Note 4) VIN = 1.1 V, TA = 25 C, L = 95 H, 3DF Coil VIN = 1.3 V, TA = 25 C, L = 95 H, 3DFCoil VIN = 1.1 V, TA = 25 C, L = 39 H, D73 Coil VIN = 1.3 V, TA = 25 C, L = 39 H, D73 Coil 0.79 V mA mA mA mA IOUT(MAX) Maximum IOUT for Converter (Notes 1,3) Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, RN ECS-TOJY106R 1K BENCH TEST CIRCUIT I(VIN) VIN LOI 300 k Note 1: Note 2: CN 10 F GND GND VOUT Note 3: IND IB VIN L = 95 H CB 10 F CS 220 pF D RS I(VOUT) IOUT Note 4: VOUT Maximum load current depends on inductor value and input voltages. Output ripple depends on filter capacitor values, ESRs and the inductor value. When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Regulation not guaranteed. CO 10 F 1K Page 6 January 1999 TOKO, Inc. TK651xx TYPICAL PERFORMANCE CHARACTERISTICS TK65130 OSCILLATOR FREQUENCY VS. TEMPERATURE 95 OUTPUT REGULATION VOLTAGE VS. TEMPERATURE 3.10 BATTERY CURRENT VS. INPUT VOLTAGE 300 250 TA = 25 C NO LOAD VOUT(REG) (V) 90 3.05 fOSC (kHz) 200 3.00 85 IB (A) 0 50 100 150 100 80 2.95 50 75 -50 0 50 100 2.90 -50 0 0 .5 1 1.5 VIN (V) 2 2.5 3 TEMPERATURE (C) TEMPERATURE (C) OUTPUT VOLTAGE VS. LOAD CURRENT 3.1 L = 95 H TOKO P/N: A682AE-014 (3DF SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 3.1 L = 100 H TOKO P/N: A636CY-101M (D73 SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 3.1 L = 39 H TOKO P/N: A636CY-390M (D73 SERIES) TA = 25 C TA = 25 C TA = 25 C 3.0 3.0 VOUT (V) VIN = 0.9 V 2.0 V 1.6 V 1.1 V 3.0 VIN = 0.9 V VOUT (V) 1.3 V 1.3 V 2.9 VIN = 0.9 V 1.6 V 2.5 V 2.8 1.1 V 2.0 V VOUT (V) 1.3 V 2.9 2.9 2.0 V 1.1 V 2.5 V 1.6 V 2.8 2.5 V 2.8 2.7 1 10 IOUT (mA) 100 2.7 1 10 IOUT (mA) 100 2.7 1 10 IOUT (mA) 100 EFFICIENCY VS. LOAD CURRENT 90 85 80 EFF (%) L = 95 F Toko P/N: A682AE-014 (3DF SERIES) SMALL COIL EFFICIENCY VS. LOAD CURRENT 90 85 L = 100 F TA Toko P/N: 636CY-101M (D73 SERIES) LARGER COIL MAXIMUM OUTPUT CURRENT VS. INDUCTOR VALUE (H) 50 40 IOUT(MAX) (mA) TA = 25 C 1.6 V EFF (%) = 25 C 1.6 V 1.3 V 80 2.0 V 75 70 1.1 V 2.0 V NO PULSE SKIPPING MODE TA = 25 C 1.3 V 1.1 V 30 20 1.1 V 10 VIN = 0.9 V 0 1.3 V 75 70 65 60 2.5 V VIN = 0.9 V 2.5 V VIN = 0.9 V 0.1 1 10 IOUT (mA) 100 65 60 0.1 1 10 IOUT (mA) 100 0 40 80 120 160 INDUCTOR VALUE (H) January 1999 TOKO, Inc. Page 7 TK651xx TK65133 ELECTRICAL CHARACTERISTICS Over operating temperature range and supply voltage range, unless otherwise specified. SYMBOL VIN IB(Q) I(VIN) I(VOUT) fOSC /T VOUT(REG) VOUT /T VOUT(LOI) VOUT(LOI) RSW(ON) EFF VULV PARAMETER Supply Voltage No Load Battery Current (Note 3) Quiescent Current into VIN Pin Quiescent Current into VOUT Pin Temperature Stability of Oscillator Regulation Threshold of VOUT Temperature Stability of VOUT(REG) VOUT During LOI Transition VOUT(LOI) Threshold Hysteresis On-resistance of SW Pin Converter Efficiency (Notes 2,3) Undervoltage Lockout Voltage TEST CONDITIONS MIN 0.90 TY P MAX 2.50 UNITS V A A A %/ C VIN = 1.3 V, IOUT = 0 mA, T A = 25 C VIN = 1.3 V, IOUT = 0 mA, T A = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, No Pulse Skipping TA = 25 C VIN = 1.3 V, IOUT = 0 mA, VIN = 1.3 V, TA = 25 C TA = 25 C VOUT 2.4 V VIN = 1.3 V, IOUT = 6 mA, L = 95 H, 3DF Coil TA = 25 C, (Note 4) VIN = 1.1 V, TA = 25 C, L = 95 H, 3DF Coil VIN = 1.3 V, TA = 25 C, L = 95 H, 3DFCoil VIN = 1.1 V, TA = 25 C, L = 39 H, D73 Coil VIN = 1.3 V, TA = 25 C, L = 39 H, D73 Coil 4 6 2.68 3.13 88 20 24 0.1 3.30 100 2.82 60 1. 0 80 0.45 6.5 8.0 12.5 20.0 134 35 40 3.40 V ppm/ C 2.93 V mV % 0.79 V mA mA mA mA IOUT(MAX) Maximum IOUT for Converter (Notes 1,3) Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, RN ECS-TOJY106R 1K BENCH TEST CIRCUIT I(VIN) VIN LOI 300 k Note 1: Note 2: CN 10 F GND GND VOUT Note 3: IND IB VIN L = 95 H CB 10 F CS 220 pF D RS I(VOUT) IOUT Note 4: VOUT Maximum load current depends on inductor value and input voltages. Output ripple depends on filter capacitor values, ESRs and the inductor value. When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Regulation not guaranteed. CO 10 F 1K Page 8 January 1999 TOKO, Inc. TK651xx TYPICAL PERFORMANCE CHARACTERISTICS TK65133 OSCILLATOR FREQUENCY VS. TEMPERATURE 95 3.40 OUTPUT REGULATION VOLTAGE VS. TEMPERATURE BATTERY CURRENT VS. INPUT VOLTAGE 300 250 TA = 25 C NO LOAD VOUT(REG) (V) 90 fOSC (kHz) 3.35 200 3.30 85 IB (A) 150 100 80 3.25 50 75 -50 3.20 -50 0 0 50 100 TEMPERATURE (C) 0 50 100 0 .5 1 1.5 VIN (V) 2 2.5 3 TEMPERATURE (C) OUTPUT VOLTAGE VS. LOAD CURRENT 3.4 L = 95 H TOKO P/N: A682AE-014 (3DF SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 3.4 L = 100 H TOKO P/N: A636CY-101M (D73 SERIES) OUTPUT VOLTAGE VS. LOAD CURRENT 3.4 L = 39 H TOKO P/N: A636CY-390M (D73 SERIES) TA = 25 C TA = 25 C TA = 25 C 3.3 VOUT (V) 3.3 3.3 VIN = 0.9 V 2.0 V 1.6 V 1.1 V VOUT (V) 3.2 VIN = 0.9 V 2.0 V 1.6 V 1.1 V VOUT (V) 1.3 V 1.3 V 3.2 1.3 V 3.2 VIN = 0.9 V 1.1 V 3.1 1.6 V 2.5 V 2.0 V 3.1 2.5 V 3.1 2.5 V 3.0 1 10 IOUT (mA) 100 3.0 1 10 IOUT (mA) 100 3.0 1 10 IOUT (mA) 100 EFFICIENCY VS. LOAD CURRENT 90 85 80 L = 95 F Toko P/N: A682AE-014 (3DF SERIES) SMALL COIL EFFICIENCY VS. LOAD CURRENT 90 85 L = 100 F Toko P/N: 636CY-101M (D73 SERIES) LARGER COIL MAXIMUM OUTPUT CURRENT VS. INDUCTOR VALUE (H) 20 16 NO PULSE SKIPPING MODE TA = 25 C TA = 25 C TA = 25 C IOUT(MAX) (mA) 1.3 V 2.0 V 2.5 V 1.6 V 2.0 V EFF (%) 75 70 1.1 V 2.5 V 1.6 V EFF (%) 1.3 V 80 75 70 1.1 V 12 8 4 1.3 V 1.1 V VIN = 0.9 V 65 60 0.1 VIN = 0.9 V 1 10 IOUT (mA) 100 65 VIN = 0.9 V 60 0.1 1 10 IOUT (mA) 100 0 0 40 80 120 160 INDUCTOR VALUE (H) January 1999 TOKO, Inc. Page 9 TK651xx SINGLE-CELL APPLICATION The TK651xx is a boost converter control IC with the power MOSFET switch built into the device. It operates from a single battery cell and steps up the output voltage to a regulated 2.7, 3.0 and 3.3 V. The device operates at a fixed nominal clock frequency of 83 kHz. In its simplest form, a boost power converter using the TK651xx requires only three external components: an inductor, a diode, and a capacitor. The analysis is easier to follow when referencing the simple boost circuit below. IPEAK di/dt = - (VOUT + Vf - VIN)/ L di/dt = VIN / L VOUT + filtering component values (consult the "Ripple and Noise Considerations" section) can be determined if needed or desired. The TK651xx runs with a fixed oscillator frequency, and it regulates by applying or skipping pulses to the internal power switch. This regulation method is called Pulse Burst Modulation (PBM). ANALYSIS OF SWITCHING CYCLE VIN LOI GND GND VOUT SW FIGURE 1: SIMPLE BOOST CONVERTER t (on) t (off) t (deadtime) THEORY OF OPERATION The converter operates with one terminal of an inductor connected to the DC input and the other terminal connected to the switch pin of the IC. When the switch is turned on, the inductor current ramps up. When the switch is turned off (or "lets go" of the inductor), the voltage flies up as the inductor seeks out a path for its current. A diode, also connected to the switching node, provides a path of conduction for the inductor current to the boost converter's output capacitor. The TK651xx monitors the voltage of the output capacitor and has a 2.7, 3.0 and 3.3 V threshold at which the converter switching becomes deactivated. So the output capacitor charges up to 2.7, 3.0 and 3.3 V and regulates there, provided that no more current is drawn from the output than the inductor can provide. The primary task, then, in designing a boost converter with the TK651xx is to determine the inductor value (and its peak current rating to prevent inductor core saturation problems) which will provide the amount of current needed to guarantee that the output voltage will be able to maintain regulation up to a specified maximum load current. Secondary necessary tasks also include choosing the diode and the output capacitor. Then the snubber and Page 10 Above is the input or inductor current waveform over a switching cycle. From an oscillator standpoint, the switching cycle consists of only an on-time and an off-time. But from an inductor current standpoint, the switching cycle breaks down into three important sections: on-time, off-time, and deadtime. The on-time of the switch and the inductor current are synonymous. During the on-time, the inductor current increases. During the off-time, the inductor current decreases as it flows into the output. When the inductor current reaches zero, that marks the end of the inductor current off-time. For the rest of the cycle, the inductor current remains at zero. Since no energy is being either stored or delivered, that remaining time is called "deadtime." This mode of the inductor current decaying to zero every cycle is called "discontinuous mode." In summary, energy is stored in the inductor during on-time, delivered to the output during off-time, and remains at zero during deadtime. January 1999 TOKO, Inc. TK651xx SINGLE-CELL APPLICATION (CONT.) The output current of the boost converter comes from the second half of the input current triangle waveform (averaged over the period or multiplied by the frequency) given by the equation: IOUT = [IPK x t(off)] x f / 2 and: (2) IPK = (VIN / L) x t(on) = VIN D / f L and: t(off) = IPK / [(VOUT + VF - VIN) / L] =(VIN D / f L) / [(VOUT + VF - VIN) / L = VIN D / f (VOUT + VF - VIN) therefore: L(MIN) = IOUT = (VIN)2 (D)2 / 2 f L (VOUT + VF - VIN) (3) which derives Equation 1 of the next section. INDUCTOR SELECTION It is under the condition of lowest input voltage that the boost converter output current capability is the lowest for a given inductance value. Three other significant parameters with worst-case values for calculating the inductor value are: highest switching frequency, lowest duty ratio (of the switch on-time to the total switching period), and highest diode forward voltage. Other parameters which can affect the required inductor value, but for simplicity will not be considered in this first analysis are: the series resistance of the DC input source (i.e., the battery), the series resistance of the internal switch, the series resistance of the inductor itself, ESR of the output capacitor, input and output filter losses, and snubber power loss. The converter reaches maximum output current capability when the switch runs at the oscillator frequency, without pulses being skipped. The output current of the boost converter is then given by the equation: IOUT = (1) (VIN)2 (D)2 2 f L (VOUT + VF - VIN) where "VF(MAX)" is best approximated by the diode forward voltage at about two-thirds of the peak diode current value. The peak diode current is the same as the peak input current, the peak switch current, and the peak inductor current. The formula is: IPK = (4) Some reiteration is implied because "L" is a function of "VF" which is a function of "IPK" which, in turn, is a function of "L". The best way into this loop is to first approximate "VF", determine "L", determine "IPK", and then determine a new "VF". Then, if necessary, reiterate. When selecting the actual inductor, it is necessary to make sure that peak current rating of the inductor (i.e., the current which causes the core to saturate) is greater than the maximum peak current the inductor will encounter. To determine the maximum peak current, use Equation 4 again, but use maximum values for "VIN" and "D", and minimum values for "f " and "L". It may also be necessary when selecting the inductor to check the rms current rating of the inductor. Whereas peak current rating is determined by core saturation, rms current Page 11 VIN D The implication from Equation 2 is that the inductor will operate in discontinuous mode. Using worst-case conditions, the inductor value can be determined by simply transforming the above equation in terms of "L": VIN(MIN)2 D(MIN)2 2 f(MAX) IOUT(MAX) [VOUT(MIN) + VF(MAX) - VIN(MIN)] VIN (VOUT + VF)(1 - D) where "VIN" is the input voltage, "D" is the on-time duty ratio of the switch, "f " is the switching (oscillator) frequency, "L" is the inductor value, "VOUT" is the output voltage, and "VF" is the diode forward voltage. It is important to note that Equation 1 makes the assumption stated in Equation 2: fL January 1999 TOKO, Inc. TK651xx SINGLE-CELL APPLICATION (CONT.) rating is determined by wire size and power dissipation in the wire resistance. The inductor rms current is given by: IPK f L D+ V OUT + VF - VIN 3 current pulses delivered from the inductor while holding a relatively smooth voltage for the converter load. Typically, the ripple voltage cannot be made smooth enough by this capacitor alone, so an output filter is used. In any case, to minimize the dissipation required by the output filter, the output capacitor should still be chosen with consideration to smoothing the voltage ripple. This implies that its Equivalent Series Resistance (ESR) should be low. This usually means choosing a larger size than the smallest available for a given capacitance. To determine the peak ripple voltage on the output capacitor for a single switching cycle, multiply the ESR by the peak current which was calculated in Equation 4. ESR can be a strong function of temperature, being worst-case when cold. The capacitance should be capable of integrating a current pulse with little ripple. Typically, if a capacitor is chosen with reasonably low ESR, and if the capacitor is the right type of capacitor for the application (typically aluminum electrolytic or tantalum), then the capacitance will be sufficient. ESR and printed circuit board layout have strong influence on RF interference levels. Special care must be taken to optimize PCB layout and component placement. THE BENEFITS OF INPUT FILTERING In practice, it may be that the peak current (calculated in Equation 4) flowing out of the battery and into the converter will cause a substantial input ripple voltage dropped across the resistance inside the battery. This becomes a more likely case for cold temperature (when battery series resistance is higher), higher load rating converters (whose inductors must draw higher peak currents), and when the battery is undersized for the peak current application. While the simple analysis used a parameter "VIN" to represent the converter input voltage in the equations, one may not know what "VIN" value to use if it is delivered by a battery that allows high ripple to occur. For example, assume that the converter draws a peak current of 100 mA for a 1 V input, and assume that the input is powered by a partially discharged AAA battery which might have a series resistance of 2 Ohms at 0 C. (Environmentally clean, so called "green" batteries tend to have higher source resistance than their "unclean" predecessors). If such partially discharged battery voltage is 1 V without load, the converter battery voltage will sag to about 0.8 V during the on-time. This can cause two problems: 1) with the effective input voltage to the converter reduced in this way, the converter output current capability will decrease, January 1999 TOKO, Inc. IL(RMS) = IPK (5) where "IPK" is the same maximized value that was just used to check against inductor peak current rating, and the term in the numerator within the radical that is added to the [on-time] duty ratio, "D", is the off-time duty ratio. Toko America, Inc. can offer a miniature matched magnetic solution in a wide range of inductor values and sizes to accommodate varying power level requirements. The following series of Toko inductors work especially well with the TK651xx : 10RF, 12RF, 3DF, D73, and D75. The 5CA series can be used for isolated-output applications, although such design objectives are not considered here. OTHER CONVERTER COMPONENTS In choosing a diode, parameters worthy of consideration are: forward voltage, reverse leakage, and capacitance. The biggest efficiency loss in the converter is due to the diode forward voltage. A Schottky diode is typically chosen to minimize this loss. Possible choices for Schottky diodes are: LL103A from ITT MELF case; 1N5017 from Motorola (through hole case); MBR0530 from Motorola (surface mount) or 15QS02L from Nihon EC (surface mount). Reverse leakage current is generally higher in Schottkys than in pin-junction diodes. If the converter spends a good deal of the battery lifetime operating at very light load (i.e., the system under power is frequently in a standby mode), then the reverse leakage current could become a substantial fraction of the entire average load current, thus degrading battery life. So don't dramatically oversize the Schottky diode if this is the case. Diode capacitance isn't likely to make much of an undesirable contribution to switching loss at this relatively low switching frequency. It can, however, increase the snubber (look in the "Ripple and Noise Considerations" section) dissipation requirement. The output capacitor, the capacitor connected from the diode cathode to ground, has the function of averaging the Page 12 TK651xx SINGLE-CELL APPLICATION (CONT.) 2) if the same battery is powering the TK651xx at the VIN pin (i.e., the normal case), then the IC may become inoperable due to insufficient VIN. This is why the application test circuit features an RC filter into the VIN pin. The current draw is very small, so the voltage drop across this filter resistor is negligible. The filter serves to average out the input ripple caused by the battery resistance. Note that this filter is optional, and the net effect of its use is the extension of battery life by allowing the battery to be discharged more deeply. A more power-efficient method comes at the price of a large capacitor. This can be placed in parallel with the battery to help channel the converter current pulses away from the battery. The capacitor must have low ESR compared to the battery resistance in order to accomplish this effectively. Still another solution is to filter the DC input with an LC filter. However, it is more likely that the filter will be either too large or too lossy. It is of questionable benefit to smooth the input if the DC loss through the filter is large. Assuming that input ripple voltage at the battery terminal and converter input is large, and that we filter the VIN pin of the IC as in the test circuit, then the parameter "VIN" in the previous equations is not usable, and we will need to use parameters to represent both the source voltage and the source resistance. SWITCH ON-RESISTANCE, INDUCTOR WINDING RESISTANCE, AND CAPACITANCE ESR The on-resistance of the TK651xx's internal switch is about 1 Ohm maximum. Using the previously stated example of 100 mA peak current, the voltage drop across the switch would reach 100 mV during the on-time. This subtracts from the voltage which is impressed across the inductor to store energy during the on-time. As a result, less energy is delivered to the output during the off-time. IB at a diode drop above the output voltage. However, the ESR of the output capacitor can increase the voltage drop across the inductor by the additional voltage dropped across the ESR when the peak current flows in it. For example, the voltage across a capacitor with an ESR of 2 Ohms (not unusual at cold temperature) would jump by 200 mV when 100 mA peak current began to flow in it. This extra voltage drop would cause the inductor current to ramp down more quickly, thus depleting the available output current. Possible choices for low ESR capacitors are: Panasonic TE series (surface mount); AVX TPS series (surface mount); Matsuo 267 series (surface mount); Sanyo OS-CON series. LOI FEATURES The Low Output Indicator (LOI) output can provide a reset signal to a microprocessor or other external system controller. When the output voltage falls below the LOI threshold (during start-up of the converter or under a current overload fault condition), the LOI signal is asserted low, indicating that the system controller (i.e., microprocessor) should be in a reset mode. This method of reset control can be used to prevent improper system operation which might occur at low supply voltage levels. The LOI threshold voltage is between 87% and 93% of the regulated output voltage value. The LOI threshold also has about 45 mV hysteresis between its on-off trigger levels. RIPPLE AND NOISE CONSIDERATIONS The filtered test circuit of the TK651xx is shown below in Figure 2. RN 1K CN 10 F VIN LOI 300 k GND GND VOUT SW ROF L = 95 F CS 220 pF D RS + 15 CU 10 F + IOUT VOUT CD 10 F If the winding resistance of the inductor increases to 1 Ohm or greater, the voltage drop across the winding resistance also subtracts from the voltage used to store energy in the core. Thus, efficiency degradation occurs. As the inductor delivers energy into the output capacitor during the off-time, its current decays at a rate proportional to the voltage drop across it. The idealized equations assume that the voltage at the switching node is clamped January 1999 TOKO, Inc. VIN 1K FIGURE 2: FILTERED TEST CIRCUIT Page 13 TK651xx SINGLE-CELL APPLICATION (CONT.) Compared to the simple boost circuit, this Filtered Test Circuit adds the following circuitry: the RC filter into the VIN pin, the RC snubber, the RC filter at the converter output, and the pull-up resistor to the LOI pin. The RC filter at the VIN pin is used only to prevent the ripple voltage at the battery terminals from prematurely causing undervoltage lockout of the IC. This is only needed when the inductor value is relatively small and the battery resistance is relatively high and the VIN range must extend as low as possible. The snubber (optional) is composed of a series RC network from the switch pin to ground (or to the output or input if preferred). Its function is to dampen the resonant LC circuit which rings during the inductor current deadtime. When the current flowing in the inductor through the output diode decays to zero, the parasitic capacitance at the switch pin from the switch, the diode, and the inductor winding has energy which rings back into the inductor, flowing back into the battery. If there is no snubbing, it is feasible that the switch pin voltage could ring below ground. Although the IC is well protected against latch-up, this ringing may be undesirable due to radiated noise. To be effective, the snubber capacitor should be large (e.g., 5 ~ 20 times) in comparison to the parasitic capacitance. If it is unnecessarily large, it dissipates extra energy every time the converter switches. The resistor of the snubber should be chosen such that it drops a substantial voltage as the ringing parasitic capacitance attempts to pull the snubber capacitor along for the ride. If the resistor is too small (e.g., zero), the snubber capacitance just adds to the ringing energy. If the resistor is too large (e.g., infinite), it effectively disengages the snubber capacitor from fighting the ringing. The RC filter at the converter output attenuates the conducted noise; the converter may not require this. Finally, the pull-up resistors at the LOI pin are needed only if this output signal is used. Most of this circuitry which appears in the test circuit has been added to minimize ripple and noise effects. But when this is not critical, the circuit can be minimized. When any DC-DC converter is used to convert power in RF circuits (e.g., pagers) the spectral noise generated by the converter, whether conducted or radiated, is of concern. The oscillator of the TK651xx has been trimmed and stabilized to 83 +/- 4 kHz with the intention of greatly minimizing interference at the common IF frequency of 455 kHz. In comparison with conventional IC solutions, where the oscillator frequency is not controlled tightly, the TK651xx can achieve as much as 20-30 dB improvements in RF interference reduction by means of its accurately controlled oscillator frequency. This IF frequency is halfway between the fifth and sixth harmonics of the oscillator. The fifth harmonic of the maximum oscillator frequency and the sixth harmonic of the minimum oscillator frequency still leave a 39 kHz band centered around 455 kHz within which a fundamental harmonic of the oscillator will not fall. Since the TK651xx operates by Pulse Burst Modulation (PBM), the switching pattern can be a subharmonic of the oscillator frequency. The simplest example, and the one to be avoided the most, is that of the converter causing every other oscillator pulse to be skipped. This means that the switching pattern would have a fundamental frequency of one-half the oscillator frequency, or 41.5 kHz. This is the eleventh harmonic, which lands at 456.5 kHz, right in the IF band. Fortunately, the energy is rather weak at the eleventh harmonic. Even more fortunate is the ease with which that regulation mode is avoided. The internal regulator comparator has a finite hysteresis. When an additional output filter is used (e.g., the RC filter of the test circuit, or an LC filter), the ripple at the regulation node is minimized. This limits the rate at which the oscillator can be gated. In practice, this means that rather than exhibiting a switching pattern of skipping every other oscillator pulse, it would be more likely to exhibit a switching pattern of three or four pulses followed by the same number of pulses skipped. Although this also tends to increase the output ripple, it is low frequency and has low magnitude (e.g., 10 kHz and 10 mV) which tends to be of little consequence. Page 14 January 1999 TOKO, Inc. TK651xx SINGLE-CELL APPLICATION (CONT.) HIGHER-ORDER DESIGN EQUATION VBB2 D IOUT = D D ( 2 L )[1- 2 L (R S + RL + RSW) ] 2 CS [VBB2+ (VOUT+ VF)2 + (VOUT + VF - VBB)2 ] D D VOUT + ROFIOUT(TGT) + (VBBRU) + VF - VBB 1 (R + RL) 2 L S 2 L ( ) - 2(VOUT + VF) The equation above was developed as a closed form approximation. In order to allow for a closed form, the design variable that requires the least approximation was "IOUT," as opposed to "L." The approximations made in the equation development have the primary consequence that error is introduced as resistive losses become relatively large. As it is normally a practical design goal to ensure that resistive losses will be relatively small, this seems acceptable. The variables used are: IOUT VOUT VBB f RS RSW RU Output current capability Output voltage Battery voltage, unloaded Oscillator frequency Source resistance (battery + filter) Switch on-state resistance ESR of upstream output capacitor IOUT(TGT) VF D L RL ROF CS Targeted output current capability Diode forward voltage Oscillating duty ratio of main switch Inductance value Inductor winding resistance Output filter resistance Snubber capacitance Deriving a design solution with this equation is necessarily an iterative process. Use worst-case tolerances as described for inductor selection, using different values for "L" to approximately achieve an "IOUT" equal to the targeted value. Then, fine tune the parasitic values as needed and, if necessary, readjust "L" again and reiterate the process. January 1999 TOKO, Inc. Page 15 TK651xx DUAL-CELL APPLICATION There are special considerations involved in designing a converter with the TK651xx for use with two battery cells. With two battery cells, the TK651xx can provide substantially more output current than a single cell input for the same efficiency. The concern is the possibility of saturating the inductor. For a single-cell input, it was only necessary to choose the current capability in accordance with the maximum peak current that could be calculated using Equation 4. For a two-cell input, the peak current is not so readily determined because the inductor can go into continuous mode. When this happens, the increase of current during the on-time remains more or less the same (i.e., approximately equal to the peak current as calculated using Equation 4, but the inductor current doesn't start from zero. It starts from where it had decayed to during the previous off-time. There is no deadtime associated with a single switching period when in continuous mode because the inductor current never decays to zero within one cycle. The cause for continuous mode operation is readily seen by noting that the rate of current increases in the inductor during the on-time is faster than the rate of decay than during the off-time. This is because there is more voltage applied across the switching during the on-time (two battery cells) than during the off-time (3 volts plus a diode minus two cells). That situation, in conjunction with a switch duty ratio of about 50%, implies that the current can't fall as much as it can rise during a cycle. So, when a switching cycle begins with zero current in the inductor, it ends with current still flowing. Continuous mode operation implies that the inductor value no longer restricts the output current capability. With discontinuous mode operation, it is necessary to choose a lower inductor value to achieve a higher output current rating (Equation 6 specifically shows "IOUT" as a function of "L"). This also implies higher ripple current from the battery. In continuous mode operation, one can choose a larger inductor value intentionally if it is desirable to minimize ripple current. The catch is that high inductance and high current rating together generally imply higher inductance size. But generally, this unrestricted inductor value allows more freedom in the converter design. The dual cell input and the continuous current rating imply that the peak current in the inductor will be at least twice as high as it would be for a single-cell input using the same inductor value. The Toko D73 and D75 series inductors are partially suited for the higher output current capability of the dual-cell configuration. For operation at a fixed maximum load, the inductor can be kept free of saturation by choosing its peak current rating equal to the converter output current rating plus the single cycle ripple current peak given Equation 4. With that guideline followed, the risk of saturation becomes only a dynamic problem. Under the situation of placing a dynamic load on the output of the converter, saturation may occur. Fortunately, unlike off-line powered converters, battery powered converters tend to be quite forgiving of dynamic saturation, due to the limitation of available power. Start-up of the converter is an example of a practically unavoidable dynamic load change (complicated by an output operating point change) that can cause saturation of the inductor. However, this particular phenomenon applies to single-cell powered converters, too. Hence, saturation is not entirely avoidable, yet does not cause system problems. It is beyond the scope of this application note to quantify the practical limitations of allowed dynamic saturation and how stressful it may be to the various components involved. It is left to the user to examine empirically the dynamic saturation phenomenon and determine what performance is acceptable. In most cases, no problem will be exhibited. Page 16 January 1999 TOKO, Inc. TK651xx STEP-DOWN CONVERTER APPLICATION HOW TO MAKE A STEP-DOWN CONVERTER USING THE TK651xx AND AN IRF7524D1 "FETKY" PART The TK651xx can be used as a controller in a step-down converter with the following two additional changes. See Fig 3. U2 IRF7524D1 VBATT R3 1k 4 1,2 R7 1k U1 TK65130 3 R4 150 SW 2 1 VOUT 4 LOI 6 5 GND R5 300 k R6 300 k + C1 220 pF C2 47 F + C3 47 F 3 5,6, 7,8 L1 VOUT 10 H R1 10 k Note: L = 10 H Toko P/N: 636CY-100M D73C Coil VIN R2 3.9 k GND FIGURE 3: STEP-DOWN CONVERTER USING THE TK651xx SCHEMATIC 1) Change the main switch orientation for use in a step-down converter. An external P-channel power MOSFET is used as the main switch in a step-down converter configuration. The gate of FET is turned on through a resistor divider being pulled down to GND by the internal output transistor of the TK651xx. This application requires both a logic level P-channel MOSFET and a Schottky diode. An IRF7524D1 "FETKY" part contains both in a small micro 8 package. 2) Change the voltage seen at the VIN pin of the TK651xx to below the regulation voltage at the VOUT pin. A resistor divider between the converter VIN and the chip VIN pin drops the voltage seen at the VIN pin. If the VIN pin is a higher voltage than the VOUT pin, the TK651xx will not regulate the output, but will continue to pulse its output transistor. WHERE TO USE THIS STEP-DOWN CONVERTER The TK65130 is a Pulse Burst Modulation (PBM) controller with a fixed duty cycle of approximately 50%. Therefore, only if VBATT is more than twice the voltage of VOUT can the converter run in CCM (continuous current mode). The converter can and does regulate in DCM (discontinuous current mode) for lighter output current loads with VIN less than twice the voltage of VOUT. But DCM produces more peak current and more ripple current than CCM. Below is a table giving some examples of where this type of step-down converter might be used. Type Battery Li-ion NiMH NiMH # of Cells 2 (Note 1) 4 (Note 2) 6 (Note 2) VBATT Range 5.4 to 8.4 V 4.4 to 5.2 V 6.6 to 7.8 V VOUT 3.0 V 3.0 V 3.0 V Typ. Max. IOUT 500 mA 500 mA 500 mA Oper. Mode DCM DCM CCM Inductor 10 H 10 H 120 H Note 1: Li-ion cell voltage range 2.7 V to 4.2 V Note 2: NiMH cell voltage range 1.1 V to 1.3 V January 1999 TOKO, Inc. Page 17 TK651xx STEP-DOWN CONVERTER APPLICATION (CONT.) THE AMOUNT OF BOARD SPACE NEEDED TO IMPLEMENT THIS STEP-DOWN CONVERTER An evaluation board for this converter has been made using a TOKO 3DF, D73 or D75 series inductor, using only 0.96 sq. inches of board space. The artwork for the surface-mount circuit board is shown below in Figure 4. Actual Size VOUT G LOI G VIN LBI 1.2 " Note: Short pin 2 to 5 for use with TK651xx FIGURE 4: TK651xx STEP-DOWN CONVERTER EVALUATION BOARD ARTWORK Page 18 .8 " January 1999 TOKO, Inc. TK651xx PULSED LOAD APPLICATION Often in the world of power conversion, the current draw of the load circuit is not constant, but rather pulsed. It is common in power supply design to size the power path large enough, and make the feedback loop fast enough to support these pulsed maximum currents. For applications where the pulse width is long or unpredictable, this approach may be warranted. However, in applications where the pulse width and maximum frequency of occurrence is predictable, such as digital cell phones or two-way pagers, it may be easier and wiser to increase the energy storage in the output filter of the power supply and keep the power path small. This leads to the need for a very large value output capacitor. Panasonic makes a series AL gold cap "super cap" which is a low voltage, large value capacitor in the one farad range. Before designing a low power DC-DC converter with a "super cap" in its output filter, it is necessary to know the loading profile (the waveform of the current going into the load from the output of the converter) of the application in which it is to be used. The converter can then be designed so that the "super cap" can be recharged in the time before the next big discharge current pulse comes along. Figure 5 is an example "super cap" charge/discharge diagram showing that the charge into the cap needs to equal the charge leaving the cap during discharge. This diagram comes from the loading and unloading profile information. In reality, some extra charge needs to go into the cap to make up for the losses caused by ESR of the cap. 1A 60 ms Note: Equal charge into and out of "supercap" 2 s (30 mA) = 60 ms (1A) Drawing not to scale IOUT 30 mA time 2s FIGURE 5: "SUPER CAP" CHARGE/DISCHARGE DIAGRAM Figure 6 is a schematic for this "super cap" example application. RN 1k CN 10 F VIN LOI GND GND VOUT CD + 10 F IOUT VOUT - IND IB VIN L = 39 F CS 220 pF RS 1k D + "SUPERCAP" 1F GOLD CAP FIGURE 6: PULSED LOAD "SUPER CAP" APPLICATION SCHEMATIC January 1999 TOKO, Inc. Page 19 TK651xx PACKAGE OUTLINE Marking Information Marking 27M 30M 33M SOT-23L-6 TK65127 TK65130 TK65133 0.6 6 5 4 Marking 1.0 1 e 2 e 0.95 3 0.32 0.95 +0.15 - 0.05 0.1 M e 0.95 e 0.95 Recommended Mount Pad 3.5 +0.3 - 0.1 2.2 0.3 (3.4) 1.4 max e1 3.0 1.2 +0.15 - 0.05 15 0 - 0.1 0.4 + 0.3 0.1 0.15 3.3 Dimensions are shown in millimeters Tolerance: x.x = 0.2 mm (unless otherwise specified) Toko America, Inc. Headquarters 1250 Feehanville Drive, Mount Prospect, Illinois 60056 Tel: (847) 297-0070 Fax: (847) 699-7864 TOKO AMERICA REGIONAL OFFICES Midwest Regional Office Toko America, Inc. 1250 Feehanville Drive Mount Prospect, IL 60056 Tel: (847) 297-0070 Fax: (847) 699-7864 Western Regional Office Toko America, Inc. 2480 North First Street , Suite 260 San Jose, CA 95131 Tel: (408) 432-8281 Fax: (408) 943-9790 Eastern Regional Office Toko America, Inc. 107 Mill Plain Road Danbury, CT 06811 Tel: (203) 748-6871 Fax: (203) 797-1223 Semiconductor Technical Support Toko Design Center 4755 Forge Road Colorado Springs, CO 80907 Tel: (719) 528-2200 Fax: (719) 528-2375 Visit our Internet site at http://www.tokoam.com The information furnished by TOKO, Inc. is believed to be accurate and reliable. However, TOKO reserves the right to make changes or improvements in the design, specification or manufacture of its products without further notice. TOKO does not assume any liability arising from the application or use of any product or circuit described herein, nor for any infringements of patents or other rights of third parties which may result from the use of its products. No license is granted by implication or otherwise under any patent or patent rights of TOKO, Inc. Page 20 (c) 1999 Toko, Inc. All Rights Reserved IC-xxx-TK651xx 0798O0.0K max January 1999 TOKO, Inc. Printed in the USA |
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